Memory cell with known state on power up

ABSTRACT

A five transistor memory cell that can be reliably read and written from a single data line. The cell includes two inverters and a pass transistor. The cell read/write circuitry includes an address supply voltage source which is maintained at a first level during write and at a second level during read, selected to reduce read disturbance. The memory cell read circuitry includes a circuit for precharging the cell data line prior to reading. The state of the memory cell is continuously available at output nodes to control other circuitry even during the read operation. Selective doping of the pull-up transistors of the inverters in the memory cell controls the initial state of the memory cell after the memory cell is powered up.

This is a continuation of application Ser. No. 07/544,071 filed Jun. 26, 1990, now abandoned, which is a continuation of application Ser. No. 07/328,840, filed Mar. 17, 1989, now abandoned, which is a divisional of application Ser. No. 07/201,509, filed Jun. 2, 1988 now abandoned, which is a continuation in part of application Ser. No. 06/777,670 filed Sep. 19, 1985 now U.S. Pat. No. 4,750,155.

FIELD OF THE INVENTION

This invention relates to a static memory cell and in particular to a five-transistor memory cell which can be reliably read and written.

BACKGROUND

FIG. 4 shows a prior art 6 transistor CMOS memory cell similar to the Intel 5101 cell. Transistors T'₁, T'₂, T'₃ and T'₄ constitute a cross-coupled latch that typically draws a steady state current of approximately one nanoampere. Transistors T'₅ and T'₆ are gating devices (pass transistors) that couple the bit lines (data lines) to the latch when the voltage on the row select line (address line) is high (5 volts). The output signal Q is a logical 1 when N channel enhancement mode transistor T'₃ is off and P channel enhancement mode transistor T'₄ is on, and it is a logical zero when these states are reversed. Reading and writing are accomplished through the left and right bit lines. For example to read the data out of the memory cell in FIG. 4, a high signal is applied to the row select, turning on transistors T'₅ and T'₆. If a logical 0 (0 volts) is on node A and a logical 1 (5 volts) is on node B, the left bit line is charged to a lower level than the right bit line. These two bit lines are typically connected to a differential amplifier (not shown) that amplifies the difference in voltage levels on the bit lines. The amplified difference is then interpreted as a logical 0 or a logical 1, according to some design convention.

To write a bit into the memory cell, the row select line is brought high (to 5 volts) and the left and right bit lines are charged to opposite states by the write driver (not shown in FIG. 4), which drives node A to the same logic level as the left bit line and node B to the same logic level as the right bit line.

The six transistor memory cell requires two gating devices (pass transistors) and two bit lines to be reliably read and written. Note that the six transistor memory cell can also be implemented in NMOS. See Holt, Electronic Circuits, John Wiley and Sons, Inc., pp. 293-294 (1978) which are incorporated herein by reference.

SUMMARY OF THE INVENTION

In contrast to the prior art, the present invention describes a five transistor memory cell which can be reliably read and written from a single data line. The memory cell includes a first and a second inverter with the output of the first inverter connected to the input of the second inverter and the output of the second inverter connected to the input of the first inverter and only a single gating (pass) transistor which is connected between the input lead of the first inverter and the single bit line.

In one embodiment of the invention the memory cell also includes a first and a second output node (lead) which continuously provides the state of the memory cell to circuitry external to the memory, for example to control the gates of external pass transistors or to provide an input signal to a logic gate.

Typically a plurality of five transistor memory cells are connected to the same data line. As one feature of the invention, means are provided for increasing the rise time on the gate of the pass transistor in order to reduce the possibility of disturbing the content of the memory cell during the read operation. As another feature of the invention, the trigger point of the first inverter is selected to be more than one threshold voltage with body effect below the voltage applied to the gate of the pass transistor and channel dimensions of the pass transistor relative to channel dimensions of the N channel and P channel transistors in the second inverter are selected to insure that the memory cell can be successfully written.

In another embodiment, circuitry is provided for precharging the data line to a first selected voltage level prior to reading a stored bit in order to reduce read disturbance.

In another embodiment circuitry is provided for charging the gate of the pass transistor to a first level during the read operation and to a second level during the write operation and for precharging the data line to a third selected voltage level prior to the read operation. The voltage levels are selected to minimize read disturbance.

The invention will be more readily understood by reference to the drawings and the detailed description.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a five transistor memory cell according to the present invention.

FIG. 2 shows a circuit for precharging the data line for the memory cell shown in FIG. 1.

FIG. 3 shows an address supply voltage source and an address driver for supplying a first selected voltage level to the address line for the memory cell of FIG. 1 during the read operation and a second selected voltage level during the write operation; FIG. 3 also shows a precharge circuit for precharging the data line to a third selected voltage level prior to the read operation.

FIG. 4 shows a prior art six transistor memory cell.

FIG. 5 shows a five transistor memory cell of the present invention using NMOS technology.

DETAILED DESCRIPTION

FIG. 1 shows one embodiment of memory cell 10 of the present invention. Memory cell 10 includes N channel enhancement mode pass transistor N₃ and inverters INV₁ and INV₂.

Inverter INV₁ includes P channel enhancement mode transistor P₁ and N channel enhancement mode transistor N₁. Source 1 of transistor P₁ is connected to the positive voltage supply having magnitude V_(CC), which is typically 5 volts plus or minus 10% although other voltages may be employed for V_(CC). Drain 2 of transistor P₁ is connected to drain 4 of transistor N₁ whose source is connected to ground. Gates 3 and 6 of transistors P₁ and N₁, respectively are connected to sense node A. The common drains 2 and 4 are connected to output node B.

Inverter INV₂ includes P channel enhancement mode transistor P₂ and N channel enhancement mode transistor N₂. Source 7 of transistor P₂ is connected to the positive voltage supply having magnitude V_(CC). Drain 8 of transistor P₂ is connected to drain 10 of transistor N₂ whose source is connected to ground. Gates 9 and 12 of transistors P₂ and N₂, respectively, are connected to output node B. Drains 8 and 10 are connected to sense node A. In operation the output signal on node B is continuously available to other circuits (not shown in FIG. 1), for example, as a gate control signal for other transistors.

Data line D_(M) is connected to sense node A via pass transistor N₃. Gate 15 of transistor N₃ is controlled by the voltage signal on address line A_(N). Source/drain 13 of transistor N₃ is connected to data line D_(M) and source/drain 14 of transistor N₃ is connected to sense node A.

One advantage of memory cell 10 is that while the signal on output node B is used, typically continuously, to control other circuits (not shown in FIG. 1), the content of memory cell 10 (i.e. the signal stored on node A) can be repeatedly checked by a read operation to verify the integrity of the memory cell's data content without degrading the output signal on node B. Furthermore, if desired, for example if the complement of the signal on node B is required to control other circuits, sense node A can also be used as an output node. This is indicated by the dotted arrow in FIG. 1. The voltage level at sense node A may be somewhat degraded during the read operation.

When sense node A stores a logical 0 and it is desired to write a logical 1 to cell 10, the signal provided at source/drain 14 of transistor N₃ must be sufficient to increase the voltage on sense node A above the trigger point of inverter INV₁ despite the pulldown effect of current flowing through transistor N₂ of inverter INV₂ (the trigger point of an inverter is the voltage at which the gate (input) voltage of the inverter equals the output voltage of the inverter). Conversely, when sense node A stores a logical 1 and it is desired to write a logical 0 to sense node A, the signal provided at source/drain 14 of transistor N₃ must be sufficient to decrease the voltage on sense node A below the trigger point of inverter INV₁ despite the pullup effect of transistor P₂ of inverter INV₂.

The operation of writing data into cell 10 and the selection of the parameters for transistors N₁, N₂, N₃, P₁, and P₂ may be understood by considering the following examples.

EXAMPLE 1

Suppose that memory cell 10 stores a logical 0, i.e. the voltage on node A is 0 volts (logical 0) and that the output signal of inverter INV₁ is V_(CC) (logical 1). Suppose that a logical 1 having a voltage level of V_(CC) on data line D_(M) is to be written to node A and that transistor N₃ is turned on by applying the signal V_(CC) to gate 15. A voltage level of V_(CC) on source/drain 13 and a voltage level of V_(CC) on gate 15 results in a voltage on source/drain 14 not higher than V_(CC) -V_(TH) (N₃). V_(TH) (N₃) is the threshold voltage of transistor N₃ with body effect. Thus the trigger point of inverter INV₁, denoted by V_(TRIG) (INV₁), is selected to be less than V_(CC) -V_(TH) (N₃). This is accomplished by selecting the ratio of the ratio of the channel width to channel length of pullup transistor P₁ to the ratio of channel width to channel length of pulldown transistor N₁ of inverter INV₁ to be sufficiently small. For example, if V_(CC) equals 5 volts and the channel width and channel length of transistors N₁ and P₁ are as given in the following table:

    ______________________________________                                         Transistor   Channel Width                                                                              Channel Length                                        ______________________________________                                         P.sub.1        5 μm   2.5 μm                                             N.sub.1      9.75 μm  2.5 μm                                             ______________________________________                                    

then the trigger point of inverter INV₁ will be less than 2 volts. Having selected the channel width and channel length of transistors P₁ and N₁ so that the trigger point of inverter INV₁ is less than V_(CC) -V_(TH) (N₃) the channel dimensions of N₂ relative to the channel dimensions of N₃ are selected so that the voltage at node A rises above the trigger point TP of inverter INV₁. When transistors N₃ and N₂ are both on, they act as a voltage divider, and the voltage at sense node A is given by V_(CC) ×(R(N₂)/(R(N₂)+R(N₃))) where R(N₂) is the channel resistance provided by transistor N₂, and R(N₃) is the channel resistance provided by transistor N₃. R(N₂) is directly proportional to L(N₂)/W(N₂) and R(N₃) is directly proportional to L(N₃)/W(N₃) where L(N₂) is the channel length of transistor N₂, W(N₂ ) is the channel width of transistor N₂, L(N₃) is the channel length of transistor N₃ and W(N₃) is the channel width of transistor N₃. By appropriately choosing the channel length and channel widths, we may ensure that V_(CC) ×(R(N₂)/(R(N₂)+R(N₃))) is greater than the trigger point TP of inverter INV₁. In one embodiment, the channel length of pass transistor N₃ is 2.5 microns and the channel width is 7.5 microns. Transistor N₂ has a channel length of 4 microns and a channel width of 4 microns. In this case R(N₂)/(R(N₂)+R(N₃)) equals 0.6. Hence the voltage on node A will rise above the trigger point of inverter INV₁. Once the voltage on sense node A rises above the trigger point, V_(TRIG) (INV₁), the output signal on node B goes low and the output signal of inverter INV₂ goes high driving sense node A to the V_(CC) level.

EXAMPLE 2

In writing a logical 0 to memory cell 10, assume the voltage on data line D_(M) is 0 volts, address line A_(N) is charged to V_(CC), and a voltage signal V_(CC) (logical 1) is stored on node A. When both transistors P₂ and N₃ are on, pullup transistor P₂ in inverter INV₂ and transistor N₃ act as a voltage divider and the voltage at sense node A is given by V_(CC) (R(N₃)/(R(N₃)+R(P₂))) where R(P₂) is the channel resistance provided by transistor P₂ and R(N₃) is the channel resistance provided by transistor N₃. The channel resistance of P₂ is directly proportional to L(P₂)/W(P₂) where L(P₂) is the channel length of transistor P₂ and W(P₂) is the channel width of transistor P₂. The channel resistance of N channel transistor N₃ is directly proportional to L(N₃)/W(N₃) where L(N₃) is the channel length of transistor N₃ and W(N₃) is the channel width of transistor N₃. These channel lengths and widths are chosen so that the voltage on sense node A falls below the trigger point of inverter INV₁. In one embodiment, transistor P₂ in inverter INV₂ has a channel length of 4 microns and a channel width of 6 microns. The fraction R(N₃)/(R(N₃)+R(P₂)) equals 0.1. In this case, the voltage on sense node A will fall below the trigger point of inverter INV₁ where the channel dimensions of inverter INV₁ are specified in the above table. Once the voltage on sense node A falls below the trigger point, the output signal on node B goes high and the output signal of inverter INV₂ goes low driving sense node A to 0 volts. The above analysis assumes that the channel resistance of the pullup and pulldown transistors of the write driver (not shown) are significantly smaller (less than 10%) than the channel resistance of transistors P₂, N₂ and N₃.

It is also desirable to be able to read the data signal stored on sense node A by transmitting this signal via pass transistor N₃ to data line D_(M) without disturbing the content of the memory. The value read is the value that appears on source/drain 13 of transistor N₃. Typically data line D_(M), which may be connected to many cells similar to cell 10 of FIG. 1, has a large capacitance compared to the capacitance of sense node A. When address line A_(N) goes high to turn on pass transistor N₃ in order to read the value stored on node A, the content of the memory (the value stored on node A) may be disturbed due to charge sharing. The following techniques can be employed to reduce the danger of disturbing the memory cell during the read operation. First, one may increase the rise time of the address line A_(N) by slowing the rate of increase of the voltage of address line A_(N). Then transistor N₃ turns on more slowly, allowing memory cell 10 to react to the disturbance caused by charge sharing without changing the content of the data stored on node A. For example, if V_(CC) is stored on node A, the rise time must be sufficiently long that the voltage on node A does not fall to V_(TRIG) (INV₁) when transistor N₃ turns on. If 0 volts is stored on node A, the rise time of the signal on address A_(N) must be sufficiently long that the voltage on node A does not rise to V_(TRIG) (INV₁) when transistor N₃ turns on. A typical address rise time should be 200 ns or more. The rise time of address line A_(N) is increased by using a "weak" (small channel width to channel length ratio) pullup transistor (not shown) in the address driver.

A second technique for avoiding disturbing the content of cell 10 during the read operation is to precharge the data line D_(M) to the value V_(TRIG) (INV₁).

Assume data line D_(M) is precharged to the value V_(TRIG) (INV₁). Assume also that a read signal of magnitude V_(CC) is applied to address line A_(N). If V_(CC) (logical 1) is stored on sense node A, then pullup transistor P₂ of inverter INV₂ and pass transistor N₃ form a voltage divider network and sense node A does not fall below V_(TRIG) (INV₁). Similarly, if 0 volts (logical 0) is stored on sense node A, then sense node A does not rise above V_(TRIG) (INV₁), since in this event transistors N₂ and N₃ form a resistor divider network and data line D_(M) is precharged to V_(TRIG) (INV₁). In one embodiment, the circuit shown in FIG. 2 is used to precharge data line D_(M) to V_(TRIG) (INV₁).

The V_(TRIG) (INV₁) precharge circuit shown in FIG. 2 includes P channel enhancement mode transistor T₁, N channel enhancement mode transistor T₂, and N channel enhancement mode pass transistor T₃. As shown in FIG. 2, source 20 of transistor T₁ is connected to the positive voltage supply V_(CC). Drain 21 of transistor T₁ is connected to drain 23 of transistor T₂ whose source 24 is connected to ground. Gates 22 and 25 of transistors T₁ and T₂, respectively, are connected to the common drain connection of transistors T₁ and T₂ which also connects to drain 26 of pass transistor T₃. Source 27 of transistor T₃ is connected to data line D_(M) and gate 28 of transistor T₃ is connected to precharge signal, φ_(precharge). The "inverter" comprising transistors T₁ and T₂ is designed to have the same trigger point as inverter INV₁, shown in FIG. 1. In the precharge cycle, the precharge signal, φ_(precharge), is set to V_(CC), which turns on N channel pass transistor T₃, and data line D_(M) is precharged to a voltage level of V_(TRIG) (INV₁) (assuming V_(TRIG) (INV₁) is lower than the voltage level of φ_(precharge) minus V_(TH),T3) The precharge signal φ_(precharge) is then brought low by control circuitry (not shown), turning off pass transistor T₃ just before the address line A_(N) connected to gate 15 of pass transistor N₃ is brought high.

The third technique for avoiding disturbing the content of memory cell 10 during the read operation is to precharge data line D_(M) to V_(CC) and set the high level of address line A_(N) to the value V_(TRIG) (INV₁). These conditions are implemented using the circuitry shown in FIG. 3. Under these conditions, when the value stored in cell 10 is V_(CC) (logical 1), pass transistor N₃ remains off and the value sensed at source/drain 13 is V_(CC) (logical 1), and sense node A is undisturbed. On the other hand, when the value stored in cell 10 is 0 volts (logical 0), the highest voltage that sense node A can be charged to is V_(TRIG) (INV₁)-V_(TH) (N₃) since N₃ is cut off when the voltage on source/drain 14 reaches this value. Hence the read "0" operation has a noise margin of V_(TH) (N₃). This is the preferred technique because the memory cell 10 is guaranteed not to be disturbed by the read operation regardless of the rise time of address line A_(N), the imbalance between the capacitance on the data line D_(M) and the capacitance on the sense node A, or the ratio of channel resistance between transistor N₃ and transistor P₂ or N₂. This third technique requires that the address line A_(N) be charged to V_(CC) for a write operation and to V_(TRIG) (INV₁) during a read operation. Symbolically, ##EQU1## The address supply voltage source can be implemented as shown in FIG. 3. Address supply circuit 90 shown in FIG. 3 includes P channel enhancement mode transistor TA₁, N channel enhancement mode transistor TA₂, N channel enhancement mode transistor TA₄, and P channel enhancement mode transistor TA₃. As shown in FIG. 3, source 30 of transistor TA₁ is connected to the positive voltage supply V_(CC). Drain 31 of transistor TA₁ is connected to drain 33 of transistor TA₂ whose source 34 is connected to drain 36 of transistor TA₄ whose source 37 is connected to ground. Source 39 of P channel transistor TA₃ is connected to V_(CC) and drain 40 of transistor TA₃ is connected to gates 32 and 35 of transistors TA₁ and TA₂ and to the common drain connection of transistors TA₁ and TA₂. Gates 41 and 38 of transistors TA₃ and TA₄ are controlled by the signal on line R/W. In the write mode, a signal of 0 volts is applied to line R/W, which turns off N channel transistor TA₄. P channel transistor TA₃ then charges V_(ADDRESS) SUPPLY to V_(CC). Note that transistor TA₃ should be designed to be sufficiently large to provide the current to address driver 70 to charge up address line A_(N) in the write mode. In the read mode, V_(CC) (logical 1) is applied to line R/W. This turns P channel transistor TA₃ off and turns on N channel transistor TA₄. By appropriately choosing channel lengths and channel widths, the circuit comprising transistor TA₁, transistor TA₂ and transistor TA₄ is designed such that the voltage at the output node 45, V_(ADDRESS) SUPPLY is the same as the trigger point of inverter INV₁ shown in FIG. 1. Thus, V_(ADDRESS) SUPPLY is equal to V_(TRIG) (INV₁). Note that transistor TA₁ should be designed to be sufficiently large to provide the current to address driver 70 to charge up address line A_(N) in the read mode. In one embodiment, transistors TA₁ and TA₃ have a channel length of 2.5 microns and a channel width of 30 microns and transistors TA₂ and TA₄ have a channel length of 2.5 microns and a channel width of 108 microns.

Address driver 70 is logically a NOR gate having input lead 54 for receiving the signal address clock and input lead 55 for receiving the signal address select.

Lead 54 provides the signal address clock to gate 58 of P channel enhancement mode transistor 52 and to inverter 56 whose output signal controls gate 61 of N channel enhancement mode transistor 50. Lead 54 is also connected to gate 65 of N channel enhancement mode transistor 66.

Lead 55 provides the address select signal to gate 59 of P channel enhancement mode transistor 53 and to inverter 57 whose output signal controls gate 62 of N channel enhancement mode transistor 51. Lead 55 is also connected to gate 63 of N channel enhancement mode transistor 64.

Transistors 50, 51, 52, and 53 comprise four transmission gates forming two parallel pairs of gates with the gates of each pair connected in series. Note that when signals address clock and address select are both low (0 volts) all four transistors 50, 51, 52 and 53 are on and N channel transistors 64 and 66 are off and thus the voltage V_(ADDRESS) SUPPLY is transmitted to address line A_(N).

P channel enhancement mode transistor 80 is connected between the voltage supply V_(CC) and data line D_(M). Data line D_(M) is precharged to V_(CC) by applying a low (0 volts) φ_(precharge) signal to gate 81 on lead 82.

The third technique described above in conjunction with FIG. 3 can be modified by replacing P channel transistor 80 by an N channel enhancement mode transistor (not shown) whose gate is controlled by the signal φ_(precharge), the complement of φ_(precharge). In this embodiment, the data line is precharged to V_(CC) -V_(T) where V_(T) is the threshold voltage of the N channel transistor.

Typically a plurality of memory cells identical to cell 10 are connected to data line D_(M). FIG. 3 shows two such memory cells having address lines A_(N) connected to address driver 70 and A_(N+1) which is connected to a corresponding address driver (not shown). In another embodiment (not shown), a rectangular memory array is formed which comprises a plurality of data lines, a plurality of address lines, and a plurality of memory cells, the memory cells attached to a given one of said data lines forming a column in the array and the memory cells attached to a given one of the address lines forming a row in the rectangular array.

The above embodiments are intended to be exemplary and not limiting. For example, while the circuits described above are implemented using CMOS technology, they may also be implemented using NMOS technology.

There can be a problem in some logic arrays if a random initial memory state on power-up causes devices controlled by the memory cells in the array to become shorted together. The shorting may cause the devices controlled by the memory cells to become unprogrammable or even damaged. This is a noticeable problem for large arrays. It is therefore desirable to control the initial memory state of the memory cells on power-up.

According to a preferred embodiment of this invention, in which the two inverters INV1 and INV2 of FIG. 1 are both CMOS, each having a p-channel pull-up transistor, P1 and P2 respectively, the two p-channel transistors P1 and P2 are differentially doped to exhibit different threshold voltages. Before power-up, all nodes in the memory cell, in particular nodes A and B, will have a voltage level of 0 volts. At power-up, as the Vcc voltage level is rising from 0 volts to the supply voltage level, the p-channel transistor requiring the lowest voltage difference between the gate and source will turn on first. In the circuit of FIG. 1, according to one preferred embodiment, transistor P2 is manufactured to have a threshold voltage of -0.7 volts while transistor P1 is manufactured to have a threshold voltage of -1.6 volts. During power-up, a low voltage applied at terminal AN to gate 15 of transistor N3 causes pass transistor N3 to be off. As the Vcc voltage level passes 0.7 volts, the gate voltages on gates 3, 6, 9, and 12 remain approximately 0 volts. Therefore, transistor P2 will begin to turn on. The increasing Vcc voltage level will then be placed by p-channel transistor P2 onto node A, which will in turn apply this increasing level to gate 3 of p-channel transistor P1, thus preventing the gate to source voltage of transistor P1 from reaching the threshold voltage of -1.6 volts during power-up. Thus at the completion of power-up the memory cell is in the known state in which node A is at a logical 1 and node B is at a logical 0.

In order to achieve the opposite initial state, the threshold voltage of transistor P1 can be manufactured to be lower than that of transistor P2. It is preferred, however, that all memory cells in a given array be manufactured so that they adopt the same initial state on power-up, in order that the ratios of channel length and channel width of the pass transistor and second inverter of each cell throughout the array, and the trigger points of inverters throughout the array can be consistent throughout the array. If it is desired that some devices controlled by memory cells be initialized at a logical one and others at a logical zero, then some devices can be controlled from node A of their respective memory cells and others can be controlled from node B of their respective memory cells.

The threshold voltages can be adjusted somewhat by adjusting the channel length of the transistors rather than by varying doping level. Transistors with shorter channel length have a somewhat lower threshold voltage than transistors with a longer channel length. However, a more sensitive method, and the preferred method of controlling threshold voltage is to dope the two p-channel transistors P1 and P2 differently. According to a preferred method of forming the structure, the doping level in the channel of transistor P2 is not changed from that of the n-well in which it is formed whereas the doping level of transistor P1 is increased during a separate masking process, during which the channel of transistor P2 is masked so as not to be affected.

The same principle described above with respect to a CMOS memory cell can also be applied to the NMOS memory cell shown in FIG. 5. In FIG. 5, transistors N5 and N6 are equivalent to transistors N1 and N2 respectively of FIG. 1. N-channel transistors N7 and N8 form resistive loads for the memory cell, being controlled by a voltage Vgg applied to their gates. Transistor N7 in one embodiment is manufactured to have a threshold voltage VtN7 of 0.7 volts while transistor N8 is manufactured to have a threshold voltage VtN8 of 1.6 volts. Thus, on power-up, as voltages Vcc and Vgg rise above 0.7 volts, transistor N7 will begin to turn on while transistor N8 remains off. Thus the voltage at node B will rise to Vgg-VtN7 and this voltage will in turn be applied to the gate of transistor N6. As the voltage Vgg-VtN7 rises above the threshold voltage of transistor N6, transistor N6 begins to turn on, pulling node A to ground, and holding transistor N5 off. As the voltage Vgg rises above 1.6 volts, transistor N8 will begin to turn on. By selecting the ratio of channel width to channel length of transistor N8 to be significantly less than the ratio of channel width to channel length of transistor N6, when both transistors N6 and N8 turn on the voltage level at node A is a logical " 0". Thus on the completion of power-up, the cell of FIG. 5 is in a known state in which node B provides a logical "1" and node A provides a logical "0".

In the embodiment of FIG. 5, it is possible for Vgg to be equal to Vcc. However, because this NMOS embodiment consumes a steady state current (in the above example, when node A is at logical "0" there is steady state current from Vcc through transistors N8 and N6 to ground), it is preferred that Vgg be separately controlled so that the Vgg voltage can be lowered to increase resistance of transistors N7 and N8, and thereby save power, or so that the cell can be turned off altogether when not in use.

Many other modifications will become obvious to one of average skill in the art in light of the above disclosure and are included within the scope of the invention. 

What is claimed is:
 1. A memory circuit comprising:a first memory transistor and a second memory transistor; a first load transistor and a second load transistor; each of said memory and load transistors having a channel, a gate, a source, and a drain, said gate controlling conductivity of said channel between said source and said drain; means for connecting said drain of said first memory transistor, said drain of said first load transistor, and said gate of second memory transistor together; means for connecting said drain of said second memory transistor, said drain of said second load transistor, and said gate of said first memory transistor together; means for connecting said gates of said first and second load transistors to a first voltage supply; means for connecting said sources of said first and second load transistors to a second voltage supply; means for connecting said sources of said first and second memory transistors to a third voltage supply; wherein a threshold voltage of said first load transistor is selected to be sufficiently lower than a threshold voltage of said second load transistor that upon power-up said drain of said first memory transistor provides a logical one and said drain of said second memory transistor provides a logical zero; said channel of said first load transistor being doped less heavily than said channel of said second load transistor.
 2. A memory circuit as in claim 1 further comprising means for connecting said gate of said second memory transistor to an input signal line.
 3. A memory circuit as in claim 2 in which said means for connecting is a pass transistor having a drain connected to said gate of said second memory transistor, a source connected to said input signal line and a gate connected to a means for controlling.
 4. A memory circuit as in claim 1 further comprising means for connecting said gate of said first memory transistor to an input signal line.
 5. A memory circuit as in claim 4 in which said means for connecting is a pass transistor having a drain connected to said gate of said first memory transistor, a source connected to said input signal line and a gate connected to a means for controlling.
 6. A memory circuit as in claim 1 further comprising means for connecting said gate of said second memory transistor to an output signal line.
 7. A memory circuit as in claim 1 further comprising means for connecting said gate of said first memory transistor to an output signal line.
 8. A memory circuit as in claim 1 in which said first and second voltage supplied are the same.
 9. A memory circuit as in claim 1 in which said first voltage supply is lower than said second voltage supply.
 10. A memory circuit as in claim 1 in which a channel of said first load transistor is not doped more heavily than a well in which said channel of said first load transistor is located and a channel of said second load transistor is doped more heavily than a well in which said channel of said second load transistor is located.
 11. A memory circuit as in claim 1 in which the channel length of said first load transistor is smaller than the channel length of said second load transistor.
 12. A memory circuit as in claim 1 in which the ratio of channel width to channel length of said second load transistor is less than the ratio of channel width to channel length of said second memory transistor. 